Regulatory bodies and wireless standards organizations require radio frequency (RF) transmitters to transmit in certain predefined frequency bands. To increase the amount of information that can be transmitted in any given frequency band (i.e., to increase what is referred to as “spectral efficiency”), many existing and soon-to-be deployed wireless communication technologies, such as Wideband Code Division Multiple Access (W-CDMA), High-Speed Packet Access (HSPA) and Long Term Evolution (LTE) cellular technologies, employ nonconstant-envelope signals.
In conventional quadrature-modulator-based transmitters, nonconstant-envelope signals must first be reduced before they are applied to the input of the transmitter's PA, and the PA must be biased in its linear region, in order to prevent signal peak clipping. However, when the PA is backed off in this manner, it is very inefficient at converting direct current (DC) power from the transmitter's power supply to radio frequency (RF) energy, and linearity is achieved at the expense of power efficiency.
Fortunately, the efficiency versus linearity trade-off of conventional quadrature-modulator-based transmitters can be avoided by using a polar modulation transmitter, instead. In a polar modulation transmitter, the amplitude information (i.e., the signal envelope) is temporarily removed from the nonconstant-envelope signal. With the signal envelope temporarily removed, the polar modulation transmitter's PA is allowed to operate in its nonlinear region, where it is more efficient at converting power from the transmitter's power supply into RF power than when configured to operate in its linear region. As explained in more detail below, the signal envelope is restored at the output of the PA by dynamically controlling the PA's power supply according to amplitude variations in the signal envelope.
FIG. 1 is a simplified drawing of a typical polar modulation transmitter 100. The polar modulation transmitter 100 includes a digital signal processor (DSP) 102; a Coordinate Rotation Digital Computer (CORDIC) converter 104; an amplitude modulation (AM) path including a first digital-to-analog converter (DAC) 106 and an amplitude modulator 108; a phase modulation (PM) path including a second DAC 110 and a phase modulator 112; a PA 114; and an antenna 116.
The DSP 102 operates to generate rectangular-coordinate in-phase and quadrature phase (i.e., I and Q) signals from bits in an incoming digital message to be transmitted. The I and Q signals are formatted in accordance with a predetermined modulation format, pulse-shaped to reduce signal bandwidth, and then coupled to inputs of the CORDIC converter 104. The CORDIC converter 104 converts the pulse-shaped I and Q signals into a digital amplitude component signal ρ representing the signal envelope and a digital phase component signal θ representing phase modulation.
The first and second DACs 106 and 110 convert the digital amplitude and phase component signals ρ and θ into analog amplitude and phase modulation signals, which are coupled to inputs of the amplitude modulator 108 and the phase modulator 112, respectively. The amplitude modulator 108 operates to modulate a DC power supply Vsupply according to amplitude variations in the analog amplitude modulation signal, to generate an amplitude-modulated power supply signal Vs(t), which is coupled to the power supply port of the PA 114. Meanwhile, the phase modulator 112 operates to modulate an RF carrier signal according to phase variations in the analog phase modulation signal, to generate a phase-modulated RF carrier signal RFin, which is coupled to the RF input port of the PA 114.
Because the phase-modulated RF carrier signal RFin has a constant envelope, the PA 114 can be configured to operate in its nonlinear region of operation. Typically, the PA 114 is implemented as a Class D, E or F switch-mode PA 114 operating between compressed and cut-off states (i.e., in a “compressed mode”), so that the output power of the PA 114 is directly controlled and modulated according to amplitude variations in the amplitude-modulated power supply signal Vs(t). By modulating the power supply port of the PA 114 in this manner, the amplitude modulation represented in the original digital amplitude component signal ρ is restored at the output of the PA 114, as the PA 114 amplifies the phase-modulated RF carrier signal RFin.
Operating the PA 114 in its nonlinear region, where it is most efficient, is highly desirable, particularly in battery-powered applications where power efficiency is an overriding concern. However, because it is operated in its nonlinear region, the PA 114 distorts the signals it amplifies. One type of distortion, known as amplitude modulation to amplitude modulation (AM/AM) distortion, results from the fact that the gain of the PA 114 compresses for higher values of the input control voltage, i.e., for higher amplitudes of the amplitude-modulated power supply signal Vs(t).
Another type of distortion known as amplitude modulation to phase modulation (AM/PM) distortion results from an undesirable phase modulation of the PA output signal RFout by an out-of-phase signal leaked from the RF input of the PA 114 to the RF output of the PA 114. The degree of AM/PM distortion introduced into the output signal depends on the amplitude of the amplitude-modulated power supply signal Vs(t) relative to the amplitude of the leaked signal. Generally, the larger the amplitude of the leaked signal is relative to the amplitude of the amplitude-modulated power supply signal Vs(t), the larger the amount of AM/PM distortion.
Various approaches that compensate for AM/AM and AM/PM distortion, i.e., approaches that “linearize” the output of the PA 114, have been proposed. One approach is to predistort the signals applied to the RF input port and power supply port of the PA 114, according to inverse responses of predetermined AM/AM and AM/PM responses of the PA 114. In this manner, when the phase-modulated RF carrier signal RFin is amplified by the PA 114, the “corrected” PA response more accurately tracks an ideal PA response compared to if no predistortion had been applied. This approach of compensating for AM/AM and AM/PM distortion is shown in FIGS. 2A and 2B.
Predistortion techniques can be effective at linearizing the output of the PA 114 of the polar modulation transmitter 100, and, as a consequence, allow the PA 114 to be operated efficiently in its nonlinear region in spite of the nonlinear characteristics of the PA 114. However, if the predistortion is not performed properly, out-of-band signal energy is generated, making it difficult to comply with out-of-band noise limitation standards.
Wireless communications standards often impose strict limits on the amount of out-of-band signal energy a transmitter is allowed to generate. Compliance with any given standard depends on how the standard is specified and on the particular type of technology involved. For example, in transmitters configured to operate according to the W-CDMA air interface in a Universal Mobile Telecommunications System (UMTS), compliance is determined by comparing the power spectral density (PSD) 308 of the output of the transmitter's PA to a spectral mask 302 defining the maximum allowable transmit power in adjacent channels, as illustrated in FIG. 3. Numerically, compliance is measured in terms of what is known as the adjacent channel leakage ratio (ACLR)—defined as the ratio of the integrated signal power in an adjacent channel (e.g., adjacent channels 306a or 306b in FIG. 3) to the integrated signal power in the desired or designated channel (e.g., desired or designated channel 304 in FIG. 3). Maximum ACLRs are specified for both uplink (mobile to base-station) and downlink (base-station to mobile) transmission. For example, compliance with the uplink ACLR specification requires that the signal power in the first adjacent channel (306a or 306b in FIG. 3) be at least −33 dB down from the average output power at the center frequency fc of the desired or designated channel 304.
Controlling out-of-band signal energy can be difficult in the design of any type of transmitter. It can be particularly difficult in the design of the polar modulation transmitter 100. One difficulty relates to the fact that the digital amplitude and phase component signals ρ and θ are processed separately in independent AM and PM paths. The level of processing performed on the digital amplitude component signal ρ in the AM path is usually different from the level of processing performed on the digital phase component signal θ in the PM path. The different levels of processing can detrimentally affect the accuracy of the AM/AM and AM/PM correction processes, result in an increase in out-of-band signal energy, and, therefore, make it difficult, or in some cases even impossible, to comply with out-of-band noise limitation standards.
It would be desirable, therefore, to have polar modulation transmitters and methods of operating polar modulation transmitters, which are capable of processing nonconstant-envelope modulation signals, and which provide accurate AM/AM and AM/PM correction, even in the presence of different levels of processing in the AM and PM paths of the polar modulation transmitters.